In the conditions of an unfavorable electromagnetic environment of a modern industrial city, the reception of amplitude-modulated signals from radio stations operating in the ranges of long, medium and short waves is very difficult. Sources of electromagnetic interference are powerful electrical equipment of industrial enterprises, electric vehicles, lighting devices, computers, household appliances, etc. The use of a magnetic antenna for radio reception in such conditions gives the best signal-to-noise ratio at the input of the radio receiver.
This is due to a different structure of the electromagnetic field near the radiation source, in the so-called near zone. After all, the radio transmitter of the radio station is far from the radio receiver by tens, hundreds and thousands of kilometers, and the distance to the source of electromagnetic interference often does not exceed tenths of a wavelength. In this case, the radio receiver antenna is in the near zone of the radiation of the interference source, as it approaches which the magnitude of the intensity vector of the magnetic component of the electromagnetic field of the interference grows proportionally to 1/r2 (where r is the distance to the radiation source), and the magnitude of the intensity vector of the electrical component is proportional to 1 /r3, which is faster. In other words, the closer the interference source, the more clearly the electric component prevails over the magnetic one in the electromagnetic field emitted by it, and this feature manifests itself to a greater extent with increasing wavelength. For the far zone, where r much more than (where is the wavelength), this effect is absent. Therefore, the use of a magnetic receiving antenna, which is insensitive to the electric component of the electromagnetic field, and also has directional properties that allow, by appropriate orientation in space, to additionally tune out from the radiation that interferes with radio reception, is more preferable in conditions of strong industrial and domestic electromagnetic interference.
To realize its advantage, the receiving magnetic antenna should not have pronounced properties of the electric antenna. For this:
The smallest dimensions of the magnetic antenna are obtained if it is made in the form of small inductors placed, for example, on a ferrite rod. The ferrite grade must correspond to the operating frequency range of the receiving antenna.
The second requirement is due to the fact that the interference EMF induced on the conductive elements of the magnetic antenna is co-phase. Effective suppression of the emerging common-mode interference provides a differential stage between the symmetrical magnetic antenna and the input of the radio receiver. To conduct experiments to identify one or another effect of common-mode interference on radio reception, the circuit shown in Fig.1 was assembled.
In the high frequency region (fD>100kHz), the circuit operates as a push-pull source follower, each of whose arms is loaded on its half of the primary winding of the matching high-frequency transformer T3. The output signal is taken from the secondary winding of this transformer through terminals XT6 and XT7. The high-frequency input signal from terminals XT1 and XT2 is fed to the primary winding of the high-frequency phase-inverting transformer T1, and antiphase signals from its secondary winding are fed to the gates of transistors VT1 and VT2. The midpoint of the secondary winding of the transformer T1 is connected to the ground wire through the capacitor C1 at high frequency, and through the direct current through the resistor R1 connected in parallel and the circuit L1:(T2.II). The DC operating point is set by resistor R2. A low-frequency (fC<12kHz) signal, which is co-phase for a push-pull source follower, is applied to the midpoint of the secondary winding of the transformer T1 from the terminals XT3 and XT4 through the low-frequency transformer T2. This signal simulated low frequency common mode interference. The low frequency common-mode component of the output signal can be observed at point XT5.
The choice of the source follower circuit is due to the fact that differential stages on FETs are most often used as input when using magnetic antennas, since they, having a very high input resistance, do not bypass the resonant system of the magnetic antenna and do not degrade its quality factor.
In order for the differential stage to effectively suppress the common-mode component of the input signal, it is necessary to ensure the symmetry of its arms. For this purpose, FETs selected with equals of zero-gate voltage drain current (IDSS) were used in the circuit in Fig.1. In addition, the winding of high-frequency transformers T1 and T3 was carried out on ring ferrite cores with a winding wire twisted three times.
To simulate high-frequency common-mode interference, a signal with a frequency of 3.25 MHz and a level of 500mV was applied to the terminals XT1 and XT2, and both gates of transistors VT1 and VT2 were connected to one of the extreme terminals of the secondary winding of the transformer T1. The level of the high-frequency signal at the output was 1.85 mV. The gain of the high-frequency co-phase signal was as follows:
To simulate low-frequency common-mode interference, the circuit was again brought into line with Fig. 1, the same signal was applied to terminals XT1 and XT2, and a sinusoidal signal with a frequency of 10 kHz and a level of 500mV was applied to terminals XT3 and XT4. The oscillogram of the observed signal at the gates of transistors VT1 and VT2 is shown in Fig.2.
As a result, the low-frequency common-mode signal, periodically changing the operating point of transistors VT1 and VT2, and hence their transconductance, modulated the output high-frequency signal as shown in the oscillogram in Fig.3.
The amplitude modulation index in this case depends only on the amplitude of the common-mode interference and in this case was about 5%. If the amplitude modulation index of AM-broadcasting is about 30%, it is easy to calculate the level of the audio signal at the output of the radio receiver caused by such low-frequency common-mode interference at its input:
and regardless of the carrier level of AM. It should also be noted that often radio receivers with a built-in magnetic antenna do not have any differential stage at the input to suppresse common-mode interference.
Such penetration of powerful low-frequency common-mode interference into the input of the radio receiver can make a broadcast listening completely impossible. Under conditions, for example, of an ordinary induction interference with a AC mains frequency of 50 Hz, a broadcast listening in the medium wave range is associated with a strong low-frequency buzz and crackle, which the author of these lines has repeatedly observed when using various types of radio receivers. To simply assess the possible level of low-frequency interference at the gates of FETs, it is enough to insert the coaxial cable with two plugs included in its kit into the high-resistance input of the oscilloscope (see Fig. 4, the vertical sensitivity was 0.02 V/div.).
The low-frequency common-mode interference must be dealt with at least two methods simultaneously:
The method of feeding the signal from the antenna to the high-resistance input of the differential stage is very important. Often, in pursuit of sensitivity, a direct connection of the oscillatory circuit of a magnetic antenna to the gates of field-effect transistors of a differential stage is used, without caring about the level of induced low-frequency common-mode interference. For example, the schemes presented in Fig. 5 cannot be considered successful. They are constructed in such a way that the high input impedance of the differential cascade made on FETs, on the one hand, does not shunt the oscillatory circuit of the resonant magnetic antenna, but, on the other hand, does not shunt the parasitic electrical “antenna” formed by the wires of the same inductors. The EMF of common-mode interference induced on them is not suppressed and applied to the gates of FETs, causing the parasitic modulation described above.
To suppress low-frequency common-mode interference at the high-resistance input of the differential stage, the communication circuit with the oscillatory circuit of the magnetic antenna can be constructed so as not to shunt this oscillatory circuit at the frequency of the useful signal and, at the same time, shunt the low-frequency common-mode interference by the low impedance of the communication circuit. Fig. 6 shows a schematic diagram of such a magnetic antenna.
The common-mode interference induced on the loop coils I and II of the magnetic antenna WA1 does not create a current in the loop and therefore does not penetrate the input of the differential stage. On the contrary, a useful signal, to the frequency of which the oscillatory circuit is tuned, creates an oscillatory circuit current in it. This current, multiplying the alternating magnetic field in the ferrite rod, creates an antiphase EMF at the opposite terminals of the coupling coil, which applied to the gates of the transistors of the differential stage. The low-frequency common-mode interference induced on the coupling coil is shunted by the low impedance of this coil on the low frequency.
A ferrite rod magnetic antenna made by the author according to the scheme shown in Fig. 6 for receiving radio broadcasting in the medium wave range showed itself well in work. The design of the antenna is shown in Fig.7.
The antenna made up of two oscillatory circuit coils and one coupling coil, each wound on a separate ferrite rod. Rods 100 mm long were taken for the oscillatory circuit coils, and a fragment 32 mm long was taken for the coupling coil. The rods are fixed as shown in Fig. 7 and pressed against each other by the ends. A coupling coil is placed in the center, wound 0.18 mm diameter wire twisted twice with a step of about 2-3 twists per centimeter of length. The coil wound in this way contains 24 + 24 turns. After winding, the outputs of the coupling coil are connected so that their phasing corresponds to the diagram in Fig.6. The signal from the coupling coil is fed to the input of a push-pull source follower, and from it, through a coaxial cable, to the input of the radio receiver intended for connecting the antenna and grounding. The inductance of identical coils I and II is calculated based on the required frequency range of the received signals and the tuning limits of the variable capacitor C1. For example, each of these coils contain 68 turns of 0.35 mm diameter wire. The coils are placed in the middle of their ferrite core each. In order to observe the phasing shown in Fig. 6, the coils are wound in such a way that the winding of one coil continues to wind the other. A variable capacitor C1, by which the antenna is tuned to resonance with the received signal, is with an air dielectric from a tube radio, but it will also work with a solid dielectric from a transistor one. Capacitors C2 and C3 are used for setting the high-frequency border of the frequency operating range, as well as for setting the symmetry of both parts of the antenna circuits (C1.1/C2/WA1.I) and (C1.2/C3/WA1.II).
The antenna was used with the old tube radio “Mir-154” and worked well. Orienting it in space to a minimum of interference, as a rule, it was possible to ensure that only atmospheric noise and the actual received radio station were heard in the receiver. With proper connection, the antenna can be used with other types of radio receivers, as well as with widely used stationary music centers, instead of the inefficient non-resonant loop antenna that they are equipped with for radio reception in the medium wave range.
Let us now consider a method for suppressing parasitic modulation of a useful signal by low-frequency common-mode interference in a differential cascade built on field-effect transistors. It is known that the gain of circuits with FETs in small signal mode depends on the transconductance of the FETs, which, in turn, is a function of the drain current. The action of an co-phase signal at the input of the differential stage according to the circuit in Fig.1 leads to an in-phase change in the drain currents of FETs, and hence to an in-phase change in their transconductance. Thus, the gain of the circuit is controlled by the common mode signal, which leads to the parasitic modulation shown in Fig.3. To suppress it, it is necessary that the transconductance of the FETs does not change under the action of a common-mode signal. This can be achieved by stabilizing the drain currents of field-effect transistors.
Fig. 8 shows a schematic diagram of a differential stage, in which, instead of a resistor in the source circuit, a current stabilizer based on bipolar transistors is used. The collector current of the transistor VT1 is equal to the sum of the source currents of the FETs VT2 and VT3. The required collector current of transistor VT1 is set by resistor R3. This current approximately equal to:
Since the operating point of the transistors is set by the current regulator on the D1 chip and does not depend on the supply voltage, this voltage can range from +6 V to +15 V.
The tests of the cascade with a current stabilizer for immunity to common mode signal were carried out under the same conditions as for the circuit in Fig. 1, with the same DC mode. At the same time, the gain of the high-frequency common-mode signal remained the same and amounted about to -45.4 dB. But the parasitic modulation caused by the low-frequency co-phase signal was significantly attenuated, as can be clearly seen from the oscillogram in Fig.9.
When powering the circuit in Fig.8 from a stabilized source, the scheme of it can be simplified by eliminating the current regulator on the D1 chip as shown in Fig.10. The collector current of the transistor VT1 will be approximately equal to:
where UVCC is the supply voltage of the circuit.
It is possible to significantly expand the dynamic range of the stage in terms of the common mode component of the input signal by applying a bipolar power supply as shown in Fig.11. In addition, bipolar power makes it possible to use FETs with a low gate-source cutoff voltage, while for the circuits in Fig. 8 and Fig. 10 it must be at least 3V.
In order not to select FETs according to the initial drain current and the gate-source cutoff voltage, the one case pair of FETs with identical in their parameters can be used.
The article presents the effective values of signal voltages.
Copyright © Sergii Zadorozhnyi, 2007